This invention relates to a voltage converter for converting an input voltage into at least two output voltages, comprising
an input resonant circuit which includes a primary winding of a transformer and to which the input voltage can be applied in periodically recurrent time intervals, PA1 at least two output circuits, each of which includes a secondary winding or a part of a secondary winding of the transformer and from which one of the output voltages can be derived,
a first one of the output voltages supplied by a first one of the output circuits constituting a high voltage and the other output voltage(s) constituting (a) voltage(s) which is (are) lower with respect to said high voltage, PA2 at least (one of) the other output circuit(s) supplying the other output voltage(s) comprising a rectifier arrangement for supplying a DC voltage as output voltage.
A voltage converter for converting an AC input voltage into a high DC voltage as well as further DC power supply voltages is known from German Offenlegungsschrift DE 195 29 941 A1. In this converter, the AC voltage applied to the input of the converter is applied to a rectifier arrangement whose output signal is applied to two series-arranged electronic switches. A series circuit of a capacitance, an inductance and a primary winding of a transformer is arranged in parallel with one of the electronic switches. A secondary winding of the transformer is arranged subsequent to the capacitance at the output. A control circuit is provided which controls the switching frequency of the electronic switches in such a way that there is a DC voltage drop of the desired value across the capacitance at the output. The transformer is provided with additional taps at the secondary side, from which the further DC power supply voltages can be derived via circuit arrangements for voltage stabilization.
If the first mentioned output in such a voltage converter is a high-voltage output which supplies a significantly higher DC output voltage as compared with the AC voltage applied to the input or as compared with the DC voltage derived therefrom by the rectifier arrangement and applied to the electronic switches, and if the further DC power supply voltages are implemented as low-voltage outputs, the capacitance at the output, which forms a parallel capacitance of the high-voltage output of the voltage converter, will decisively determine the operating behavior of this voltage converter due to the then selected higher transformation ratio between the primary winding and the secondary winding of the transformer, because this capacitance is transformed to a proportionally high value at exactly this high transformation ratio at the primary side of the transformer. On the other hand, parallel capacitances dimensioned in a comparable manner at the additional taps at the secondary side, constituting low-voltage outputs of the voltage converter for the comparably low further DC power supply voltages, have only a small influence at the primary side of the transformer because they are only transformed accordingly with the lower transformation ratios at the primary side.
It has been found that the parallel capacitance of the high-voltage output in a conventional implementation of such a voltage converter exerts a significant, predominant influence on the primary side of the transformer even when it is not constituted by an additional component but only by the parasitic winding capacitance of the secondary winding.
In contrast, the effect of parasitic winding capacitances from the secondary windings or the parts of the secondary winding connected to the taps for the further DC power supply voltages are negligibly small when transformed to the primary side of the transformer.
In the case of loads on the different secondary windings and taps, i.e. when electric power is derived via the high voltage or the further DC power supply voltages, there is a mutual influence in a voltage converter of the type described in the opening paragraph, which mutual influence is referred to as cross-regulation and becomes manifest in that the output voltages, i.e. the high voltage and the further DC power supply voltages are not only dependent on their mutual ratio of turns of the secondary windings or parts of the secondary winding but also on their respective loads. This means that, in spite of regulating one of the DC voltages at a fixed value, fluctuations occur in the other DC voltages which are dependent on the loads of the outputs.
It may be attempted to limit these fluctuations by maximally reducing the leakage inductances of the secondary windings or the parts of the secondary winding of the transformer. However, this solution is subject to limits, particularly because of the design of the core and the bobbin of the transformer, the requirement of mutual voltage insulation of the individual windings of the transformer and the minimum number of turns of the separate windings which, inter alia, is determined by the maximally admissible magnetic induction in the core. In practice, this has the result that, for given uses of the voltage converter described and the implementation of the transformer required for these uses, the requirements simultaneously imposed on a synchronous operation of the voltages supplied at the secondary side cannot be met. This means that the cross-regulation in such voltage converters cannot be kept within the prescribed limits so that the voltage converter of the type described is not usable for said application.
For the purpose of elucidation, FIG. 1 shows the problem of cross-regulation of a voltage converter for converting an input voltage UB into a high DC voltage and a further power supply voltage. This voltage converter comprises two series-arranged switches 4, 5 which are alternately periodically conducting and non-conducting as choppers between the input voltage UB and ground 6. The series arrangement of a resonant capacitance CRES, a resonant inductance LRES and the primary winding of a transformer T between junction points A and B is connected parallel to the second one of these switches, denoted by the reference numeral 5. The resonant capacitance CRES and the resonant inductance LRES are elements of an input resonant circuit of the voltage converter, which also includes the primary winding of the transformer T.
The transformer T has two secondary windings, a first of which is connected to junction points H and V and a second is connected to junction points J and K. The transformer T is dimensioned so that the junction points H and V supply an AC voltage of a high amplitude and the junction points J and K supply a further AC voltage. The first secondary winding between the junction points H and V is therefore also referred to as a high-voltage winding. The further AC voltage at the second secondary winding between the junction points J and K has an essentially lower amplitude; in contrast to the AC voltage of a high amplitude (high voltage) at the first secondary winding, it will hereinafter be referred to as the low voltage.
Each secondary winding of the transformer T is included in an output circuit. The first output circuit, which includes the first secondary winding between the junction points H and V, is implemented for supplying the high DC voltage, and the second output circuit, which includes the second secondary winding between the junction points J and K, is implemented for supplying the further power supply voltage. In the voltage converter shown in FIG. 1, the first output circuit comprises a voltage multiplier 7 whose output is connected to a load capacitance CL2 at which the high DC voltage is available during operation. A parallel capacitance CP which, dependent on the implementation of the voltage converter, may be constituted by the parasitic winding capacitance of the first secondary winding of the transformer T in the first output circuit or by a separate component, is inserted between the junction points H and V.
The output circuit of the voltage converter of FIG. 1, including the second secondary winding, comprises a bridge rectifier with four diodes D1, D2, D3 and D4. The output of this bridge rectifier is connected to a load capacitance CL3 at which the further power supply voltage is available during operation.
In FIG. 1, no capacitance is inserted between the junction points J and K. Due to the choice of the transformation ratio between the second secondary winding between the junction points J and K and the primary winding between the junction points A and B, on the one hand, and the transformation ratio between the first secondary winding between the junction points H and V and the primary winding, on the other hand, the influence of a parasitic winding capacitance of the second secondary winding, transformed to the primary side of the transformer T, is negligible as compared with the parallel capacitance CP in this voltage converter.
To explain the operation of the voltage converter of FIG. 1, FIG. 2 shows an equivalent circuit diagram of the transformer T. It comprises, between the junction points A and B as an equivalent circuit diagram for the primary winding, a series arrangement of a primary leakage inductance LS1 with a parallel arrangement of a principal inductance LH of the transformer T and a primary winding 1 of an ideal transformer. At the secondary side, the equivalent circuit diagram of FIG. 2 comprises, for the first secondary winding, the winding 2 of the ideal transformer in series with a leakage inductance LS2 at the secondary side between the junction points H and V, and a series arrangement of a further secondary winding 3 of the ideal transformer with the associated leakage inductance LS3 at the secondary side between the junction points J and K. The values for the leakage inductances can be unambiguously determined from the technical data of the transformer T to be used. The equivalent circuit diagram of FIG. 2 further shows the parallel capacitance CP for the first secondary winding 2 between the junction points H and V.
To elucidate the problem of cross-regulation, FIG. 3 is a further simplification of the equivalent circuit diagram of the transformer T, together with the most important other elements of the voltage converter of FIG. 1. The equivalent circuit diagram is simplified in that the elements of the transformer T at the secondary side are transformed with the associated transformation ratios to the primary side of the transformer T so that a simplified, DC-coupled equivalent circuit diagram is obtained. In this circuit diagram, elements already described have the same reference symbols. The reference LS22 denotes the leakage inductance LS2 at the secondary side, transformed to the primary side. Similarly, the reference LS33 denotes the leakage inductance LS3 at the secondary side, transformed to the primary side of the transformer T. The reference CP22 denotes the parallel capacitance CP in its form transformed to the primary side of the transformer T. Similarly, the references H22, V22, J33 and K33 denote the junction points of the transformed circuit corresponding to the junction points H, V, J and K, respectively. During operation of the voltage converter of FIG. 1, the voltage uAB is obtained between the junction points A and B for the equivalent circuit diagram of FIG. 3, the voltage uH22V22 is obtained between the junction points H22 and V22, and the voltage uJ33K33 is obtained between the junction points J33 and K33. The primary circuit of the voltage converter, i.e. the input resonant circuit with the primary winding of the transformer T, is further simplified in FIG. 3. The leakage inductance LS1 is combined with the resonant inductance LRES to a resultant resonant inductance LS1RES. As, in contrast, the principal inductance LH can usually be considered as being high-ohmic, it is not shown in FIG. 3. In operation, a current i flows in the input resonant circuit.
FIG. 4 shows the voltages in the equivalent circuit diagram of FIG. 3 for operation of the voltage converter of FIG. 1, and the current i in the input resonant circuit. The start is an instant ts when the transformed parallel capacitance CP22 is completely discharged. Similarly, the voltage uH22V22 at the instant ts is zero. At a proportionally small transformed leakage inductance LS22 and a slow variation of the current i, the voltage produced at the transformed leakage inductance LS22 may initially be ignored. In a first approximation, the voltage uAB in this operating condition thus corresponds to the voltage uH22V22. Both rise, starting from zero at the instant ts. In this operating condition, the bridge rectifier D1, D2, D3, D4 is still blocked, while no current flows in the second output circuit via the junction points J33, K33, and the voltage uJ33K33 directly follows the variation with respect to time of the voltage uAB. The current i completely flows in the parallel capacitance CP22 and charges this capacitance.
The operating condition described above changes at the instant t0 when the parallel capacitance CP22 is charged to such an extent that the voltage uAB and hence the voltage uJ33K33 reach a value U which represents the further DC power supply voltage at which the load capacitance CL3 is charged during operation, transformed to the primary side of the transformer T. At the instant t0, the bridge rectifier D1, D2, D3, D4 thus becomes conducting and the voltage uJ33K33 maintains the value U. The equivalent circuit diagram for this state of operation is shown in FIG. 5 in which elements already explained have the same reference symbols.
If the voltage produced at the transformed leakage inductance LS33 is also ignored in the equivalent circuit diagram of FIG. 5, the voltage uAB maintains the value U from the instant t0. In a voltage converter without the cross-regulation effect, the voltage uH22V22 would also maintain the value U from the instant t0. However, also after the instant t0, the voltage at the (transformed) parallel capacitance CP22 further rises. This variation is shown in FIG. 4.
To explain the rise of the voltage uH22V22, FIG. 6 shows a section of the equivalent circuit diagram of FIG. 3, in which the elements already described again have tile same reference symbols. The section shown in FIG. 6 comprises the transformed leakage inductance LS22 at the secondary side and the transformed parallel capacitance CP22. It is shown that these elements in the operating situation described constitute a series circuit of an inductance and a capacitance to which a voltage jump corresponding to the voltage U is applied at the instant t0. The voltage which is formed at the parallel capacitance CP22, transformed to the primary side, has a sinusoidal variation whose amplitude is proportional to the value of the current i in the input resonant circuit at the instant t0 and inversely proportional to the value of the transformed parallel capacitance CP22. Moreover, this amplitude is inversely proportional to the duration of the sinusoidal voltage variation which is again determined from the reciprocal value of the square root of the product of the transformed parallel capacitance CP22 and the transformed leakage inductance LS22. The voltage overshoot, i.e. the amplitude of the sinusoidal variation of the voltage uH22V22 after the instant t0, as shown in FIG. 4, is thus proportional to the current i in the input resonant circuit and hence, in accordance with the equivalent circuit diagram of FIG. 3 in the transformed, secondary leakage inductance LS22 at the instant t0 and proportional to the square root of this transformed leakage inductance LS22 and inversely proportional to the square root of the transformed parallel capacitance CP22. This overshoot results in a disturbing cross-regulation.
Since the current through the transformed leakage inductance LS22 corresponds to the current i in the input resonant circuit at the instant t0, it is determined by the power consumption of the voltage converter and its dimensioning. The dependence of the voltage overshoot on the elements of the voltage converter in the form described could lead to the expedient of increasing the value of the parallel capacitance CP or reducing the leakage inductance LS2. However, a reduction of the voltage overshoot is not achieved thereby because an increase of the parallel capacitance CP directly involves an increase of the current i at the instant t0. Moreover, narrow limits, as explained hereinbefore, are imposed on a reduction of the leakage inductance LS2. However, without a reduction of the voltage overshoot, a reduction of the cross-regulation is not possible.